• 专利附录
  • 专利说明
  • 权利要求
  • 类似技术
  • 同族专利
  • 引用文献
  • 法律状态
  • 优先权
    • 专利摘要:
    To more accurately determine the temperature of a DC link capacitor (C) of a DC link converter (1), a device and a method are described in which the DC link capacitor (C) is modeled as a series circuit of a spare capacity (CS) and a spare series resistance (ESR) in which a modeled capacitor current (iCm) flows via the equivalent series resistor (ESR). From the modeled capacitor current (iCm) and the value of the equivalent series resistance (ESR), a modeled capacitor power loss (PC) is calculated by means of a first relationship of the form = (,) from which the capacitor temperature (TC) is determined by means of a predetermined temperature model. No direct measurement of capacitor temperature (TC), capacitor current (iC) or capacitor power loss (PC) is required. For example, a measurement of the capacitor voltage (μC) and a further calculation of the modeled capacitor current iCm and finally the capacitor power loss (PC) is sufficient. The method can be used for observing and processing the capacitor temperature (TC), in particular switching off an element, preferably at least part of the intermediate circuit converter (1) when exceeding a preset maximum temperature, for example. Also, the method for determining the time profile of the capacitor temperature (TC (t)) and further determining the residual life (RL) of the intermediate circuit capacitor (C) of a predetermined relationship, preferably on the Arrhenius formula can be used.
    公开号:AT518194A1
    申请号:T50048/2016
    申请日:2016-01-29
    公开日:2017-08-15
    发明作者:Ing Dipl (Fh) Klaus Doppelhammer；Eder Johann；Ing Gerhard Mayrhofer-Huber Dipl
    申请人:Bernecker + Rainer Industrie-Elektronik Ges M B H；
    IPC主号:
    专利说明:

    Temperature of a DC link capacitor
    The subject invention relates to a method and a device for determining a temperature profile of a DC link capacitor of a DC link converter, which includes at least one n-phase inverter. Furthermore, the use of the method according to the invention for determining the remaining service life of the DC link capacitor from the temperature profile is described.
    DC link converters are used in a variety of circuits, including servo amplifiers, frequency converters, converters, regenerators, etc., particularly for electric motor drives such as stepper motors or brushless DC motors, etc. These are often capacitors, and in particular electrolytic capacitors (ELKOs) in DC link installed as energy storage elements, these DC link capacitors serve to smooth the DC link voltage. By a desired space saving can be installed in some systems only a limited number of such capacitors. The lifetime of the overall circuit or circuit parts is often primarily dependent on the life of the capacitor of the DC link. Since high performance requirements can lead to thermal problems of the DC link capacitors, it is advantageous if the remaining service life of the DC link capacitor is known. The capacitor life in turn can be determined in a known manner by means of the Arrhenius formula from a time temperatu rverlauf the capacitor. For this purpose, the ambient temperature and / or the Rippeistrombelastung the capacitor are often used. An elicitation of the capacitor life is thus usually only possible in the laboratory, since only here predetermined circumstances (ambient temperature, current waveform, etc.) can be created. This method is often used for a priori design and optimization of capacitors. For known requirements, it is then possible to use capacitors with desired properties.
    However, it is often desirable, especially with previously unknown requirements, that the (residual) service life of a capacitor can also be predicted or determined during operation, and in particular in a DC link converter.
    The KR 2013 0110553 A offers such an approach, whereby the converter supplies an electric motor in the publication. Based on the torque of the electric motor and the angular velocity of the associated rotor, the total electrical power of the converter circuit is calculated. Based on the voltage applied to the intermediate circuit capacitor and the total power determined, the ripple current is subsequently estimated. Subsequently, the course of the temperature of the DC link capacitor and, as a result, the service life of the DC link capacitor are determined from the course of the estimated ripple current on the basis of stored characteristic curves. The disadvantage here is that the determined total power of the inverter does not correspond to the power consumed at the DC link capacitor and therefore does not actually match the estimated Rip-pelstrom actually with the current through the DC link capacitor. Thus, the determined temperature profile is very inaccurate. Furthermore, additional sensors for determining the torque and the speed are required.
    One could measure the capacitor current and the capacitor voltage or the power consumed at the DC link capacitor or the capacitor temperature. However, the entire additional measurement technology required for this purpose is often undesirable in a DC link converter for reasons of cost and space. In particular, the measurement of the capacitor temperature (ie the temperature in the interior of the DC link capacitor) would require special, equipped with integrated temperature sensors and thus expensive, DC link capacitors.
    The aim of the invention is thus to determine the temperature of a DC link capacitor in a DC link circuit with less effort more accurately, in particular, the above-mentioned disadvantages should be avoided.
    This objective is achieved by modeling the link capacitor as a series circuit of equivalent capacitance and equivalent series resistance and flowing a modeled capacitor current across the equivalent series resistor. From the modeled capacitor current and the value of the equivalent series resistor, a modeled capacitor power loss is calculated by means of a first relationship of the form Pc = f (iCm, ESR), from which the capacitor temperature is determined by means of a predetermined temperature model.
    The modeled capacitor power dissipation drops in the capacitor model with the equivalent capacitance and series resistance at the series resistor. Thus, the capacitor current flowing through the series resistor and thus through the intermediate circuit capacitor is itself modeled, and on this basis the modeled capacitor power loss is calculated at the intermediate circuit capacitor. Based on this, the capacitor temperature can be determined using a known temperature model, for example a PT1 model.
    For calculating the condenser temperature according to the invention, a calculation unit can be provided.
    Advantageously, Pc = i2Cm ESR is used as the first connection.
    The modeled capacitor current can be modeled using a measured capacitor voltage and the equivalent capacitance CS, using a second relationship of the form iCm = f (uc, CS).
    Advantageously, the second relationship i
    used.
    For this purpose, a voltage measuring unit and a differentiating unit may be present, wherein the voltage measuring unit determines a capacitor voltage applied to the intermediate circuit capacitor and supplies it to the differentiating unit, and the differentiating unit determines the modeled capacitor current after multiplication by the equivalent capacitance.
    Advantageously, the modeled capacitor current is divided into a low-frequency component and a high-frequency component, wherein the low-frequency component of the modeled capacitor current is used to determine a first modeled power loss and the high-frequency component of the modeled capacitor current is used to determine a second modeled power loss. The modeled capacitor power dissipation is calculated from the sum of the first and second modeled power losses.
    The separation into a high-frequency and a low-frequency component of the modeled capacitor current may depend on the switching frequency of the inverter of the DC link converter. When PWM control is used, the PWM switching frequency can thus be used to determine the boundary between the high-frequency and low-frequency components of the modeled capacitor current.
    Advantageously, the low-frequency component of the modeled capacitor current is determined by averaging the capacitor voltage and / or a time derivative of the capacitor voltage and / or the modeled capacitor current. As a result, high-frequency components are cut off and subsequently have no influence on the first modeled power loss. When the capacitor voltage is averaged, only low frequency portions of the capacitor voltage are processed further. After the second relationship, that is to say for example by means of time derivation and multiplication with the equivalent capacitance, the low-frequency component of the modeled capacitor current is concluded. However, if only the second relationship is used, first the modeled capacitor current with high frequency and low frequency components is calculated, and then averaged to cut off the high frequency component and obtain the low frequency component. However, this requires a very high sampling frequency.
    The calculation of the low-frequency modeled capacitor current can take place in a low-frequency unit, with low-pass filters being provided before or after the differentiation unit in order to carry out the mean value filtering. Furthermore, the low-pass filter is used to avoid aliasing problems and to smooth the signal.
    The high-frequency component of the modeled capacitor current can be calculated from an inverter current caused by the inverter, which is advantageously carried out in a high-frequency unit. As an inverter current here is the intermediate-circuit side inverter current to consider. The inverter current can be measured directly or calculated from the phase-side phase currents of the converter, or the inverter, the measurement of two phase currents would be sufficient, since the third phase current can be calculated from the two first phase currents. If several inverters are present, the high-frequency component of the modeled capacitor current can be determined from the sum of the (intermediate-circuit side) inverter currents.
    Subsequently, the square of the high frequency portion of the modeled capacitor current can be calculated by finding the square of the arithmetic mean of the inverter current and the arithmetic mean of the square inverter current and subtracting the square of the arithmetic mean of the inverter current from the arithmetic mean of the square inverter current. For this purpose, a number of averaging agents may be present.
    The arithmetic mean of the inverter current may be time discrete by sampling the inverter current at a first sampling rate in a first time period, generating a first number of samples of the inverter current and dividing the sum of the samples of the inverter current by the first number.
    Similarly, the formation of the arithmetic mean of the square inverter current can be accomplished by sampling the square inverter current at a second sampling rate in a second time period producing a second number of square inverter current samples and the sum of the square inverter current samples and the second Number is divided.
    The determined capacitor temperature can be recorded over time, for example in a memory unit provided for this purpose.
    The device can be used to observe and process the capacitor temperature, in particular a shutdown of an element, preferably at least a portion of the DC link converter when exceeding a, for example, preset, maximum temperature. This can serve to avoid overheating of the capacitor or to minimize the duration of the elevated temperature.
    It is a further object of the present invention to determine the remaining service life of a capacitor of a DC link of a DC link converter. This is achieved by the method according to the invention also being used for determining the time profile of the capacitor temperature and further for determining the remaining service life of the DC link capacitor from the time profile of the capacitor temperature by means of a predetermined relationship, preferably via the Arrhenius formula.
    When using the Arrhenius formula, the time profile of the capacitor temperature is used, it can also be included the ambient temperature of the DC link capacitor, for which a temperature sensor can be used. Advantageously compared to the prior art, the capacitor current and consequently the capacitor power loss is thus modeled directly, and not only the total current of the DC link converter is calculated from the total power consumption and used for the determination of the temperature.
    The subject invention will be explained in more detail below with reference to Figures 1 to 6, which show by way of example, schematically and not by way of limitation advantageous embodiments of the invention. It shows
    Fig. 1 is a simplified circuit diagram of a DC link converter in the form of a servo amplifier
    2 shows an equivalent circuit diagram of a DC link capacitor
    3 shows a calculation unit with model block
    3a shows a thermal model of the DC link capacitor
    4 shows a general pulse pattern of an inverter with associated phase currents
    5 shows a structure of the calculation unit for calculating the capacitor temperature and the remaining service life
    6 shows a part of the calculation unit for calculating the high-frequency component of the modeled capacitor current
    As an exemplary use of a DC link converter 1, a simplified circuit diagram of a servo amplifier is shown in FIG. The DC link converter 1 is the input side to the supply network 2 and the output side connected to an electric motor 3, wherein the supply network 2, the phase currents ii ', i2', 13 "to the DC link 1 supplies and the DC link converter 1, the inverter currents i-ι, i2,13 on the electric motor 3 supplies. The supply of the DC link converter 1 could also be done by another voltage source, for example using DC link terminals.
    On the side of the supply network 1, a rectifier 4 is installed, which converts the input side m-phase AC voltage into a DC voltage that feeds a DC Zwi-schenkreis 6 (DC link). If a supply of the DC link converter takes place via another voltage source, under certain circumstances no rectifier 4 may be required. On the output side, on the side of the electric motor 3, an n-phase inverter 5 is provided, which, e.g. from a PWM control (for reasons of clarity, only shown for a circuit breaker of the inverter 5) is driven. The present invention relates to inverter 5 with an arbitrary number n of phases, by way of example an inverter with n = 3 phases is listed. In FIG. 1, the supply network 2 with m = 3 is designed in three phases, as is the rectifier 4, and the electric motor 3, which is only to be understood by way of example, since the number of phases of the supply network, the rectifier 4 and the electric motor 3 do not have to match. The rectifier 4 is configured in the present example in a known manner for each of the m phases as a half-bridge and consists of two diodes per phase. Of course, the rectifier 4 could also be present in another embodiment, for example as an active front end (AFE) of an input / regenerator. The inverter 5 is realized in a likewise known manner by circuit breakers, for example IGBTs or MOSFETs. As an energy-storing element of the DC intermediate circuit 6, an intermediate circuit capacitor C, for example, an electrolytic capacitor (ELKO), installed. Of course, the DC link capacitor C can also be designed as a combination circuit (series connection, parallel connection) of a plurality of individual DC link capacitors. The rectifier 4 supplies the rectifier current iREc and the inverter 5 draws the current i | NV as a function of the electrical load. About the DC link capacitor C, the capacitor current ic flows.
    FIG. 2 shows the equivalent circuit diagram of the intermediate circuit capacitor C used in the invention. The intermediate circuit capacitor C is modeled via an equivalent capacitance CS and a series resistance which is assumed to be in series and is generally frequency-dependent. About the equivalent resistance ESR, which is usually a few milliohms and can be assumed to be known flows the modeled capacitor current iCm.
    From the modeled capacitor current iCm and the value of the equivalent series resistor ESR, a modeled capacitor power loss Pc is calculated in a calculation unit BE in a power calculation unit 10 by means of a first predetermined relationship of the form Pc = f (ic, ESR), for example Pc = ic2ESR. According to the invention, the current capacitor temperature Tc is further determined from the modeled capacitor power loss Pc by means of a known temperature model 11 implemented in a model module M, as shown in FIG. The aim in the first step is therefore to approximate the capacitor current ic through the modeled capacitor current iCm in order to be able to calculate the modeled capacitor power loss Pc and then further the capacitor temperature Tc dependent thereon.
    3a shows a possible temperature model 11 of the intermediate circuit capacitor C, which allows the capacitor temperature Tc of the intermediate circuit capacitor C to be calculated from the modeled capacitor power loss Pc. The temperature model 11 is shown here as a model with a delay element 1st order, a so-called PT1 element, it being understood that other temperature models can also be used. In the temperature model 11, the modeled capacitor power loss Pc represents a thermal power source in the thermal network. By a thermal capacitance Cth and a thermal resistance Rth, a thermal voltage sets at the thermal capacitance Cth, which corresponds to the capacitor temperature Tc. For this purpose, as shown in FIG. 1, the ambient temperature TA of the DC link capacitor, which is represented in the temperature model 11 as a thermal voltage source, is measured by means of the temperature sensor TS. Alternatively, the ambient temperature TA could also be estimated by a suitable method, for example from a specified (maximum) ambient temperature. The ambient temperature TA and the modeled capacitor power loss Pc are fed to the temperature model 11 and from this the capacitor temperature Tc is calculated.
    The modeled capacitor current iCm can be modeled, for example, using the measured capacitor voltage uc and the equivalent capacitance CS, where a second relationship of the form ic = f (uc, CS), for example
    can be used.
    The time derivation can take place, for example, in the form of a discrete derivative over a discrete time period ΔΤ with a time index k:
    In this case, the discrete period ΔΤ thus corresponds to the inverse sampling rate and is generally located well above the inverse of the PWM switching frequency. In terms of frequencies, the sampling frequency is well below the PWM switching frequency, which would unwantedly cut off high frequency components of the modeled capacitor current icm.
    In order to directly calculate the modeled capacitor current iCm via this method, a sampling frequency as a multiple of the PWM switching frequency would therefore be required in the case of discrete execution of the derivative, thus resulting in a sampling frequency in the MHz range. Although this would be technically feasible, it would often not be economical due to the necessary components and, in addition, may cause further problems, e.g. Problems of electromagnetic compatibility (EMC).
    An advantageous embodiment of the determination of the modeled capacitor current iCm will therefore be described below with reference to FIGS. 5 and 6.
    The modeled capacitor current iCm is advantageously divided into a low-frequency component iCL and a high-frequency component iCp (also called pulse-frequency component), the low-frequency component iCi_ of the capacitor current iCm for determining a first power loss PCl and the high-frequency component iCp of the capacitor current for determining a second power loss PCp is used. The capacitor power loss Pc is calculated from the sum of the first power loss PCl and the second power loss PCp.
    Pc = Pcl + Pcp = ic2ESP = Lcl2PSPl + icp2ESRp
    Strictly speaking, the squares of the high-frequency component iCp2 and the low-frequency component iCi_2 of the capacitor current are processed. The division of the square of the capacitor current ic2 into the sum of the squares of the low-frequency component iCi_2 and the high-frequency component iCp2 is for all (also non-periodic) signals, which are divided into a DC component (mean-free) and an AC component, in the interval in which the averaging is done, possible. In the case of sinusoids, the product icL'icp is e.g. is integrated over a period and is zero due to the orthogonality of the sinusoids. Quantitatively, the first power loss PCl and the second power loss PCp are approximately equal, and the equivalent series resistance ESR is divided into the equivalent series resistances ESRL and ESRP, which in turn are known in advance and can be obtained, for example, from the data sheet of the DC link capacitor.
    A low-frequency unit BL, which is preferably arranged in the calculation unit BE, carries out the calculation of the low-frequency component of the capacitor current iCi_, by the capacitor voltage uc and / or the time derivative of the capacitor voltage
    and / or the modeled capacitor current iCm is averaged. For this purpose, as shown in FIG. 5, a low-pass filter TP may be arranged in front of a differentiation unit D in order to average-filter the capacitor voltage uc. Of course, it would also be possible (additionally) to attach a (further) low-pass filter TP after the differentiation unit D to the time derivative of the capacitor voltage
    mittelwertzufiltern. Likewise, a mean value filtering of the determined modeled capacitor current ic would be possible. It is important that the high-frequency components are cut off. The capacitor voltage uc, as indicated in FIG. 1, is measured by means of a voltmeter V and supplied to the low-frequency unit BL. The measurement of the intermediate circuit voltage, which corresponds to the capacitor voltage uc, is usually implemented in a DC link converter 1 and thus represents no additional expense. The low-frequency component of the modeled capacitor current iCL is subsequently squared and multiplied by the low-frequency equivalent resistance ESRL to the first power loss PCl to calculate.
    The cause of the low-frequency component of the modeled capacitor current iCi_ is primarily to be sought on the side of the rectifier 4, whereby even low-frequency processes such as load changes on the inverter 5 side influences. The high-frequency component of the modeled capacitor current iCp is generally caused by the high-frequency switching in the inverter 5. In the case of the application of a PWM control corresponds to the high-frequency frequency component of the PWM switching frequency and above, so mostly from 5kHz. Accordingly, frequencies below the PWM switching frequency are to be considered as a low-frequency component. Since the capacitor voltage uc is measured at the DC link capacitor C, thus basically low-frequency components of the side of the electric motor 3, and not only both sides of the rectifier 4, are taken into account.
    If a DC link converter 1 is supplied by the supply network 1 (for example, a 50 Hz three-phase network, as shown in FIG. 1), a ripple with a frequency of 300 Hz (mains ripple, 6 half-waves per period) is produced after rectification. Superimposed on this ripple are the repercussions of the electric motor 3 due to the process that is being run (process ripple). This frequency is in practice below 1000 Hz. The first power loss PCl thus consists of the mains ripple and the process ripple. An additional choke mounted in the rectifier could reduce this mains ripple, which, however, causes additional costs and requires more space. If the DC link converter 1 is supplied by a DC voltage instead of the supply network 2 by a DC link, so eliminates the mains ripple and for the first power loss Pcl only the proportion of the process ripple remains. Moreover, by operating the electric motor 3 in the so-called S1 mode (i.e., the load does not change), the first power loss PCi_of zero is obtained.
    The high-frequency component of the modeled capacitor current iCp is calculated in a high-frequency unit BP, as described with reference to FIG. 6, from the inverter current i | NV caused by the inverter 5. For this purpose, advantageously, the inverter current i! Nv produced by the inverter 5 (intermediate-circuit-side) is first calculated as the sum of the phase currents h, i 2, i 3 of the inverter. For a general pulse pattern of the upper switches S1, S2, S3 of a 3-phase inverter 5, the phase currents i-1, i2, i3 of the inverter 5 result according to the following table:
    For ease of illustration, the general pulse pattern is also shown in FIG. The inverter current iinv caused by the inverter 5 results particularly advantageously from the measured first and second phase currents h and i2, the third phase current i3 being determined by means of the first Kirchhoff rule with i3 = -u -i2, with the aid of FIG Table above and shown in Fig. 4 pulse pattern to i-iNv = Slii "T ^ 2 ^ -2 + ^ 3 (-i ^ -i2)
    Of course, all phase currents h, i2, i3 or the inverter current i! Nv could also be measured directly. The phase currents h, i2, i3 are usually measured in a DC-link converter 1 and are thus available. From the inverter current i | NV, the high-frequency component of the modeled capacitor current icp (or the square of the high-frequency component iCp2) can be calculated using the following method:
    The square of the high-frequency portion of the modeled capacitor current iCp2 is calculated by taking the arithmetic mean value iINV of the inverter current i | Nv (which corresponds to the DC component) and the arithmetic mean value of the square inverter -7 -2
    Current iINV be determined and the square of the arithmetic mean iINV is subtracted from the mean value of the square inverter current iINV2.
    The averaging of the inverter current is advantageously time-discrete. With a first sampling rate TSi, in a first time period T-1 a first number N-i of samples of the inverter current i | Nv is obtained. Further, the Ni samples of the inverter current iiNv are summed up and divided by the first number N-i.
    Analogously, a second number N2 of samples of the square-wave inverter current i! Nv2 can be generated in a time-discrete manner at a second sampling rate TS2 in a second time period T2. Further, the N-i samples of the square inverter current i | Nv2 are summed up and divided by the second number N2. the arithmetic mean of the inverter current iINV is time-discrete, in that the inverter current iiNv is sampled at a first sampling rate TSi in a first time period T-ι, generating a first number Ni of samples of the inverter current iinv and the sum of the first number Ni of samples of the inverter current iinv and divided by the first number Ni.
    In FIG. 5, these averaging takes place in two mean value images MWB located in the radio-frequency unit BP. Of course, it would also be possible for only an averaging image MWB to assume this task by being logically correctly connected to the respective branch. For arithmetic averaging, there is thus an accumulation of N samples of the inverter current i | Nv in the period T and a subsequent division by the number of recorded samples. For the determination of the arithmetic mean of the square iINV2, the square of the inverter current i | Nv2 in the period T is summed up and divided by the number N of recorded samples.
    The square of the high-frequency component of the modeled capacitor current iCp is subsequently multiplied by the high-frequency equivalent resistance ESRp in order to calculate the second power loss PCp. The first power loss PC1 and the second power loss PCp are added to obtain the capacitor power dissipation Pc.
    The temperature model 11 is implemented in FIG. 5 in a model component M to which the ambient temperature TA of the intermediate circuit capacitor C and the capacitor power loss Pc are supplied. As a result, the capacitor temperature Tc is output.
    5, the low-frequency unit BL, and the high-frequency unit BP, as well as the model module M and the power calculation unit 10 are advantageously provided in the calculation unit BE.
    As shown in Fig. 5, a residual life unit RE may be provided which processes the detected capacitor temperature Tc as the condenser temperature history Tc (t). For this purpose, the current capacitor temperatures Tc determined in each time step are recorded as a time characteristic. For this purpose, the capacitor temperature profile Tc (t) can be stored in a memory unit SE. In the residual life unit RE, an estimate of the residual life RL of the intermediate circuit capacitor C is carried out with the capacitor temperature profile Tc (t), with the aid of the known Arrhenius formula. The Arrhenius formula states that the residual life RL of the DC link capacitor C is doubled per approx. 10 ° C temperature reduction. It is therefore possible to start from the residual life RL at maximum temperature and to convert it to the current condenser temperature Tc. As an example, assume a total life of a capacitor of 1000 hours at a maximum temperature of 105 ° C. If the DC link capacitor C is always operated at 95 ° C, so it has a total life of 2000 hours. The total life, as well as the remaining life RL is always to be seen in an indication in hours with respect to an assumed capacitor temperature Tc. Therefore, it is helpful to specify the residual life RL in percent. If the above-mentioned intermediate circuit capacitor C is operated for 500 hours at a condenser temperature of 95 ° C., then a residual service life RL of 1500 hours remains at 95 ° C., giving a residual service life RL of 750 hours at 105 ° C., or alternatively a residual service life RL of 75 % corresponds. If the DC link capacitor C continues to operate for 100 hours at 105 ° C, the residual life RL is reduced to 650 hours at 105 ° C, or 1300 hours at 95 ° C, or generally to 65%. Advantageously, an action is performed after reaching a predetermined minimum residual life RL. It can be output, for example, at a residual life RL of 20%, a signal to effect a replacement of the DC link capacitor C.
    It should be expressly understood that the inventive method can also be used on DC link converter 1 with multiple inverters 5 on the output side. In this case, several inverters 5 are connected in parallel to the DC intermediate circuit 6. In this case, the high-frequency component of the modeled capacitor current iCp can be calculated analogously as described above, wherein the currents iINVi are summed over the i inverters 5 to the current i | NV. This sum is used for the further calculation of the arithmetic mean value iINV and the square of the effective value iINV2 and as a consequence of the high-frequency component of the capacitor current ICP and finally the course of the capacitor power loss Pc and the capacitor temperature Tc.
    权利要求:
    Claims (17)
    [1]
    claims
    1. A method for determining a capacitor temperature (Tc) of a DC link capacitor (C) of a DC link converter (1), the at least one n-phase inverter (5), characterized in that the intermediate circuit capacitor (C) as a series connection of a spare capacity (CS) and a Replacement Series Resistor (ESR) is modeled and via the equivalent series resistance (ESR) a modeled capacitor current (iCm) flows and that from the modeled capacitor current (iCm) and the value of the equivalent series resistance (ESR) by means of a first relationship of the form Pc = f (iCm, ESR) a modeled capacitor power loss (Pc) is calculated, from which by means of a predetermined temperature model, the capacitor temperature (Tc) is determined.
    [2]
    2. The method according to claim 1, characterized in that the first connection Pc = icm ESR is used.
    [3]
    3. Method according to claim 1 or 2, characterized in that the modeled capacitor current (iCm) is modeled using a measured capacitor voltage (uc) and the equivalent capacitance (CS), wherein a second relationship of the form iCm = f (uc, CS) is used.
    [4]
    4. The method according to claim 3, characterized in that the second context

    is used.
    [5]
    5. The method according to any one of claims 1 to 4, characterized in that the modeled capacitor current (iCm) is divided into a low-frequency component (iCi_) and a high-frequency component (iCp), wherein the low-frequency component of the modulated capacitor current (icL) for determining a first modeled power loss (PCl) and the high-frequency portion of the modeled capacitor current (iCp) is used to determine a second modeled power loss (Pcp) and the modeled capacitor power loss (Pc) from the sum of the first modeled power loss (PCl) and the second modeled power loss (Pcp) is calculated.
    [6]
    6. The method according to claim 5, characterized in that the low-frequency component of the modeled capacitor current (iCi_) is determined by the capacitor voltage (uc) and / or a time derivative of the capacitor voltage (^ uc) and / or the modeled capacitor current (iCm ) is averaged.
    [7]
    7. The method according to claim 5 or 6, characterized in that the high-frequency component of the modeled capacitor current (iCp) is calculated from an inverter current (i! Nv) caused by the inverter (5).
    [8]
    A method according to claim 7, characterized in that the square of the high frequency portion of the modeled capacitor current (iCp2) is calculated by finding the -2 square of the arithmetic mean of the inverter current (iINV) and the arithmetic mean of the square inverter current (iINV2) and subtracting -2 square of the arithmetic mean of the inverter current (iINV) from the arithmetic mean of the square inverter current (iINV2).
    [9]
    9. The method according to claim 8, characterized in that the formation of the arithmetic mean value of the inverter current (iINV) is time-discrete by the inverter current (iiNv) is sampled at a first sampling rate (TSi) in a first period (Ti), wherein a first Number (Ni) of samples of the inverter current i | Nv) is generated and the sum of the samples of the inverter current (i | Nv) and divided by the first number (Ni).
    [10]
    A method according to claim 8, characterized in that the arithmetic mean of the square inverter current (iINV2) is time discretely sampled by sampling the square inverter current (iinv2) at a second sampling rate (TS2) in a second time period (T2) a second number (N2) of samples of the square inverter current (iinv2) is generated and the sum of the samples of the square inverter current (iinv2) and divided by the second number (N2).
    [11]
    11. Device for determining the capacitor temperature (Tc) of a DC link capacitor (C) of a DC link (6) of a DC link converter (1), characterized in that a calculation unit (BE) is present, the DC link capacitor (C) as a series connection of a spare capacity (CS ) and a spare series resistance (ESR), wherein a modeled capacitor current (iCm) flows via the equivalent series resistance (ESR) and from the modeled capacitor current (iCm) and the value of the equivalent series resistance (ESR) by means of a relationship of the form Pc = f (iCm, ESR) calculates a modeled capacitor power loss (Pc) and determines therefrom the capacitor temperature (Tc) by means of a predetermined temperature model (M).
    [12]
    12. The device according to claim 11, characterized in that a differentiating unit (D) and a voltage measuring unit (V) are present, wherein the voltage measuring unit (V) determines a voltage applied to the intermediate circuit capacitor (C) capacitor voltage (uc) and the differentiating unit (D) supplies and the differentiating unit (D) determines the modeled capacitor current (iCm).
    [13]
    13. The device according to claim 11, characterized in that, a low-frequency unit (BL) is provided, which is configured to calculate a low-frequency component of the modeled capacitor current (iCi_) by before and / or after the differentiating unit (D) a number Low-pass filter (TP) are provided.
    [14]
    14. Device according to one of claims 11 to 13, characterized in that a high-frequency unit (BP) is present, which calculates a high-frequency portion of the modeled capacitor current (iCp) from an inverter current (iinv) caused by the inverter (5).
    [15]
    15. The device according to claim 14, characterized in that a number of averaging (MWB) is present, which calculate an average value of the inverter current (iINV), or an average value of the square inverter current (iINV2).
    [16]
    16. Use of a method according to one of claims 1 to 9 for observing and processing the capacitor temperature (Tc), in particular switching off an element of the DC link converter (1) when a maximum temperature is exceeded.
    [17]
    17. Use of a method according to one of claims 1 to 9 for determining the time profile of the capacitor temperature (Tc (t)) and determining the remaining life (RL) of the intermediate circuit capacitor (C) from the time profile of the condenser temperature (Tc (t)) by means a predetermined relationship, preferably on the Arrhenius formula.
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    同族专利:
    公开号 | 公开日
    DK3199929T3|2019-06-03|
    EP3199929A1|2017-08-02|
    US20170219441A1|2017-08-03|
    EP3199929B1|2019-04-17|
    US10416028B2|2019-09-17|
    AT518194B1|2020-10-15|
    CA2956510A1|2017-07-29|
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    DE102004035723A1|2004-07-23|2006-02-16|Siemens Ag|Residual lifetime determination method for electrolytic capacitor in frequency converter, determines core temperature and power loss to calculate aging speed|
    JP2006229130A|2005-02-21|2006-08-31|Kansai Electric Power Co Inc:The|Method of estimating and managing lifetime of power unit|
    DE202014101916U1|2013-04-29|2014-05-12|Sma Solar Technology Ag|Inverter for feeding electrical power into a power grid and photovoltaic system|EP3836336A1|2019-12-10|2021-06-16|Wobben Properties GmbH|Device and method for angular stabilization of a virtual synchronous machine|US20040264216A1|2003-06-25|2004-12-30|Alexander Mednik|Switching power converter and method of controlling output voltage thereof using predictive sensing of magnetic flux|
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    EP2637030A1|2012-03-05|2013-09-11|ebm-papst Mulfingen GmbH & Co. KG|Method for life-cycle monitoring of an electrolyte condenser and device with a monitored electrolyte condenser|
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    DE102013203299A1|2013-02-27|2014-08-28|Robert Bosch Gmbh|Method for determining equivalent series resistance of direct current link circuit capacitor in inverter of motor vehicle, involves determining equivalent series resistance, based on ratio of determined voltage jump and determined current|EP3477314B1|2017-10-24|2020-09-30|Mitsubishi Electric R & D Centre Europe B.V.|A method for on-line monitoring a dc-bus capacitor|
    CN109813462B|2018-12-11|2020-09-04|深圳市法拉第电驱动有限公司|Capacitor core temperature processing method, device, equipment and computer readable storage medium|
    DE102019117369A1|2019-06-27|2020-12-31|Ebm-Papst Mulfingen Gmbh & Co. Kg|Circuit and method for monitoring an intermediate circuit capacitor|
    EP3883113A1|2020-03-20|2021-09-22|Mitsubishi Electric R & D Centre Europe B.V.|A method for on-line monitoring a dc-bus capacitor|
    法律状态:
    优先权:
    申请号 | 申请日 | 专利标题
    ATA50048/2016A|AT518194B1|2016-01-29|2016-01-29|Method and device for determining the temperature of an intermediate circuit capacitor|ATA50048/2016A| AT518194B1|2016-01-29|2016-01-29|Method and device for determining the temperature of an intermediate circuit capacitor|
    EP17152999.3A| EP3199929B1|2016-01-29|2017-01-25|Method and device to measure temperature of an intermediate circuit capacitor|
    DK17152999.3T| DK3199929T3|2016-01-29|2017-01-25|METHOD AND ESTABLISHMENT TO MEASURE A TEMPERATURE PROCEDURE FOR AN INTERMEDIATE CONDENSER|
    US15/418,048| US10416028B2|2016-01-29|2017-01-27|Temperature of a link capacitor|
    CA2956510A| CA2956510A1|2016-01-29|2017-01-27|Temperature of a link capacitor|
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